Project C08 - Real-Time Multiband THz Passive Imaging in Silicon Technologies
Systems
Principal Investigator: Prof. Dr. Ullrich Pfeiffer
Achieved results and methods
Previous challenging research questions have been addressed in the 2nd phase of C08 with a focus on researching the fundamental limitations of lens- and antenna-coupled, silicon-integrated THz direct detectors and fully implemented total power radiometers with broadband LNA pre-amplification.
Silicon-integrated THz Direct Detectors
To push the sensitivity of antenna-coupled THz direct detectors towards 1, the focus was not only set on the design of new detector structures based on an optimum detector antenna-co design. Further, an analysis of the fundamental limit of the THz rectification process in advanced SiGe HBT technologies was performed, as detectors implemented during the 1st phase of C08 have already shown an excellent performance in terms of NEP with values around 1.9 at around 300 GHz. In the 2nd phase, the fundamental influence of the device parasitics, namely Rb, Re, Rc, Cbe, Cbc was investigated on the maximum THz rectification and the potential for further improvement estimated,
based on the development of a nonlinear equivalent device model [10]. The primary outcome is that due to further antenna optimization, the optimum device NEP cannot be improved by more than 20 % and THz rectification is possible in the low-power saturation region with only a factor of two performance degradation in terms of NEP [11]. Detector improvements are limited to the device’s internal parasitics, mainly on the low-pass networks formed by internal Cbe, Re and Cbe, Rb, as well as the intrinsic device nonlinearity which can only be changed with access to the process technology. Nevertheless, to enhance the sensitivity towards passive imaging, new dual-polarized antenna structures have been implemented in an advanced SiGe HBT technology with ft/fmax of 470/650 GHz, benefitting from the simultaneous measurement of two polarizations, increasing the overall receiver thermal sensitivity by a factor of two. The developed direct detectors were fully packaged with a 3 mm hyper-hemispherical silicon lens and characterized in a free-space measurement setup with the Friis transmission equation, including the measured detector directivity. The performance of one of these detector circuits reaches a minimum NEP of 2.3 at 300 GHz in 1 polarization path and in theory, 1.63 if both polarization paths are connected. The dual-polarized antenna was further used to demonstrate polarization-sensitive imaging [12]. In summary, an equivalent NEP of 4.7 in an excellent equivalent rectangular bandwidth of 512 GHz is achieved, theoretically (Eqn. 1) leading to a minimum NETD of 0.67 K in a 1 Hz bandwidth. As shown in Fig. 1, the investigated detector shows the lowest minimum NEP and highest bandwidth across the current state-of-the-art for THz direct detectors implemented in silicon. Finally, the resolution limits of silicon-integrated THz cameras have been investigated [13] and a 32 x 32 light-field camera SoC was implemented that currently lacks the sensitivity for passive imaging [14, 15].
Passive Imaging with THz Direct Detectors
The detector NETD was determined with a black body source under optical chopping at 1.5 kHz, as shown in Fig. 2. First, the detector's thermal responsivity Rv(T) was measured as output voltage slope vs. black body temperature. This resulted in a detector thermal responsivity of up to 23.2 nV/K for a single polarization and 44.38 nV/K measured with both polarizations. By calculating the output noise voltage and thermal responsivity ratio, the corresponding NETD in a 1 Hz readout bandwidth approached 0.64 K, well correlating to the values de-embedded from the CW (NEP) measurements. With these results, the presented THz direct detector is the first detector room temperature operated THz direct detector with an NETD below 1 K with 500 ms integration time.
Broadband LNA
Based on the previously described analysis of the optimum detector sensitivity, the only way of enabling real-time passive imaging with sub 100 ms integration time in a direct detection scheme is the introduction of an LNA in front of the detector. Due to the challenging requirements for passive imaging stated in Eq. 1, at least 30 dB of gain and 100 GHz bandwidth are necessary to achieve a real-time NETD of 50 mK with a NF<10 dB. Achieving such a high gain and bandwidth with a center frequency above 200 GHz (enhancing the system’s imaging resolution) is extremely challenging. Most common LNA designs in the literature are based on classical transmission-line matching networks. Here, a high gain over 100 GHz bandwidth has not been reported in silicon technologies before the 2nd C08 phase. During the current project phase, the design of a broadband, five-stage differential cascode LNA shown in Fig. 3 a) was implemented and fully characterized with the help of broadband on-chip baluns [16]. With an accurate EM transistor core modeling, its optimization and the development of novel broadband distributed, coupled-line matching transformers, an unprecedented 3-dB bandwidth of 146 GHz, a maximum gain of 34.6 dB and a minimum NF of 8.4 centered around 204 GHz [17] was achieved, as shown in Fig. 3 b). An excellent correlation between measurements and simulations was reported. In total, a state-of-the art gain-bandwidth product of 421 THz was reported, only at the cost of a moderate power consumption of 152 mW. The presented LNA shows the best performance among the state-of-the-art of silicon-integrated LNAs with a center frequency above 200 GHz.
Dual-Polarized Radiometer
The most challenging total power radiometer design aspects are maintaining the LNA gain of more than 30 dB, bandwidth of more than 100 GHz and low NF when implemented with an antenna and a direct detector. A bandwidth above 100 GHz is of special importance as it improves the overall system NETD and counteracts the antenna implementation losses as well as the increasing NF. Another optimization aspect is the coverage of a second orthogonal linear polarization path, which improves the total radiometer NETD by a factor of at least √2. In total, the implemented radiometer consists of a dual-polarized antenna, a 5-stage LNA with several modifications in each polarization path and a differential common-base detector circuit at the LNA outputs. The most critical design parameters are the radiometer input and output matching networks. For output matching, two different output matching transformers were designed in both polarization paths, realizing a slight frequency tuning. The radiometer, shown in Fig. 3 a), was packaged with a 9 mm hyper-hemispherical silicon lens and characterized in a free-space measurement and a black body setup as shown in Fig. 2 a). In total, a minimum NEP of 23 fW/√Hz and a maximum responsivity of 59 MV/W were measured in the free-space measurement setup with a corresponding equivalent RF bandwidth of 150 GHz indicated in Fig. 4 b) centered around 200 GHz in a 1 Hz noise bandwidth [18]. This is the first silicon-integrated total power radiometer operating above 200 GHz with more than 100 GHz bandwidth and a thermal responsivity of up to 60 μV/K and a minimum NETD of 23 mK shown in Fig. 4.
Passive Imaging with a Dual-Polarized Total Power Radiometer
To finally demonstrate real-time passive imaging, a human finger (TFinger ~ 35°C) was recorded by motorized raster scanning and a chopping frequency of 300 Hz. The radiometer output signal was directly coupled to an external 16-bit ADC, where the chopped signal was read out at each scanning position after a fast Fourier Transformation (FFT) with a sampling rate of 2 MS/s. 131072 samples were captured per data point, resulting in an overall frame rate of 15 frames per second. After applying a flat-top window function, the signal amplitude was further read out, resulting in a 57 Hz equivalent noise bandwidth. The most important thing to note is the maximum linear SNR of 26.5 in the THz image of the uncovered finger. The maximum signal amplitude of 720 µV with the chopping factor de-embedded further corresponds to a measured temperature difference of roughly 13.2 K to the ambient room temperature, when divided by the thermal radiometer sensitivity for the used detector bias point (55 μV/K, Vbe = 780 mV, Vce = 1 V). To demonstrate the inherent see-through ability of THz waves and highlight their advantage among conventional IR cameras, additional images of the human finger with cardbox covering have been recorded with a FLIR T450sc IR camera operating in a spectral range from 40 to 23 THz (7.5 to 13 µm). As shown in Fig. 5, with cardbox covering, the amplitude of the THz images reduces only by 30 %, while it completely blocks the IR radiation. In summary, the radiometer's excellent suitability for a passive detection of concealed objects in a real-time environment is demonstrated.
Antenna-Coupled Microbolometer
By implementing antenna-coupled microbolometers and silicon-integrated electronic circuits together it was planned to extend the THz detection capability towards the IR frequency range in one imager. This can be achieved by a co-integration and coupling silicon-integrated antenna incorporating the THz direct detectors to the microbolometer implementation. Different integration concepts were analyzed to thermally isolate the antenna from the microbolometers, and capacitive coupling has been chosen. In this case, the electronic detectors can directly act as a lid for the microbolometer vacuum package. Nevertheless, microbolometers have to be modified to provide the second capacitor electrode lying on the top metal layer, which in IR applications is used as absorber. To enhance the power transfer, the capacitance was optimized by precisely changing the distance between the antenna and microbolometer and the horizontal alignment. However, typical process variations are in the range of 5 µm (vertical) and 2 µm between the lid's edge and the bolometer's center. The horizontal alignment showed insufficient deviations up to 20 µm.
By optimizing process parameters, the precision of the horizontal alignment was improved to < 2 µm limited by the toolset. For vertical distance control, additional support pillars of solder material inside the imager array were added, preventing bending due to ambient pressure, as shown in Fig. 6. With a pixel pitch of 17 µm in the bolometer array and the antennas being larger than 100 µm, 1 out of 36 microbolometers was combined with the antenna. Several microbolometers were used as mechanical support without reducing the sensor function as shown in Fig. 7. This way, a technology for capacitively coupled antennas and microbolometers was realized.
Selected project-related publications